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19-1850; Rev 0; 10/00
Digital Camera Step-Down Power Supply
General Description
The MAX1802 provides a complete power-supply solution for digital still cameras and video cameras by integrating two high-efficiency step-down DC-DC converters and three auxiliary step-up controllers. This complete solution is targeted for applications that use either three to four alkaline cells or two lithium-ion (Li+) cells. The main step-down DC-DC controller accepts inputs from 2.5V to 11V and regulates a resistor-adjustable output from 2.7V to 5.5V. It uses a synchronous rectifier to regulate the output with up to 94% efficiency. An adjustable operating frequency (up to 1MHz) facilitates designs for optimum size, cost, and efficiency. The core step-down DC-DC converter accepts inputs from 2.7V to 5.5V and regulates a resistor-adjustable output from 1.25V to 5.5V. It delivers 500mA with up to 94% efficiency. The three auxiliary step-up controllers can be used to power the digital camera's CCD, LCD, and backlight. The MAX1802 also features expandability by supplying power, an oscillator signal, and a reference to the MAX1801, a low-cost slave DC-DC controller that supports step-up, single-ended primary inductance converter (SEPIC), and fly-back configurations. The MAX1802 is available in a space-saving 32-pin TQFP package (5mm x 5mm body), and the MAX1801 is available in an 8-pin SOT-23 package. An evaluation kit (MAX1802EVKIT) featuring both devices is available to expedite designs. o 2.5V to 11V Input Voltage Range o Main DC-DC Controller 94% Efficiency +2.7V to +5.5V Adjustable Output Voltage Up to 100% Duty Cycle Independent Shutdown o Core DC-DC Converter 94% Efficiency Up to 500mA Load Efficiency Output Voltage Adjustable Down to 1.25V Independent Shutdown o Three Auxiliary DC-DC Controllers Adjustable Maximum Duty Cycle Independent Shutdown o Power, Oscillator, and Reference Outputs to Drive External Slave Controllers (MAX1801) o Up to 1MHz Switching Frequency o 3A Supply Current in Shutdown Mode o Internal Soft-Start o Overload Protection for All DC-DC Converters o Compact 32-Pin TQFP Package
Features
MAX1802
________________________Applications
Digital Still Cameras Digital Video Cameras Hand-Held Devices Internet Access Tablets PDAs DVD Players
Ordering Information
PART MAX1802EHJ TEMP. RANGE -40C to +85C PIN-PACKAGE 32 TQFP
Note: Refer to the separate data sheet for MAX1801EKA in an 8pin SOT.
Typical Operating Circuit
MAIN INPUT 2.5V TO 11V CORE MAX1802 MASTER CCD CCFL TFT OSC POWER MAX1801 SLAVE REF MOTOR
Pin Configuration appears at end of data sheet.
________________________________________________________________ Maxim Integrated Products
1
For price, delivery, and to place orders, please contact Maxim Distribution at 1-888-629-4642, or visit Maxim's website at www.maxim-ic.com.
Digital Camera Step-Down Power Supply MAX1802
ABSOLUTE MAXIMUM RATINGS
VDDM, VH, ONM to GND .......................................-0.3V to +12V PGNDM, PGND to GND ........................................-0.3V to +0.3V VH to VDDM .............................................................-6V to +0.3V VL to VDDM ............................................................-12V to +0.3V VL, ONC, ON1, FB_, DCON_ to GND ......................-0.3V to +6V VDDC, REF, OSC, COMP_ to GND ..............-0.3V to (VL + 0.3V) DHM, DLM to PGNDM............................-0.3V to (VDDM + 0.3V) LXM to PGNDM ......................................-0.6V to (VDDM + 0.6V) DL1, DL2, DL3, LXC to PGND ................-0.3V to (VDDC + 0.3V) Continuous Power Dissipation (TA = +70C) 32-Pin TQFP (derate 11.1mW/C above +70C)........889mW Operating Temperature Range ...........................-40C to +85C Junction Temperature ......................................................+150C Storage Temperature Range. ............................-65C to +150C Lead Temperature (soldering, 10s) .................................+300C
Stresses beyond those listed under "Absolute Maximum Ratings" may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability.
ELECTRICAL CHARACTERISTICS
(Circuit of Figure 1, VVDDM = 6V, VVDDC = 3V, PGNDM = PGND = GND, DCON1 = REF, VONM = 3V, VONC = VON1 = VDCON2 = VDCON3 = 0, TA = 0C to +85C, unless otherwise noted. Typical values are at TA = +25C.)
PARAMETER GENERAL Input Voltage Range SUPPLY CURRENT Shutdown Supply Current (from VDDM and VDDC) Main DC-DC Converter Supply Current (from VDDM) Main DC-DC Converter Supply Current (from VDDC) Main plus Core Supply Current (from VDDC) Main plus Auxiliary 1 Supply Current (from VDDC) Main plus Auxiliary 2 Supply Current (from VDDC) Main plus Auxiliary 3 Supply Current (from VDDC) Total Supply Current (from VDDC) VL REGULATOR VL Output Voltage VL Supply Rejection VL Undervoltage Lockout Threshold VL Switchover Voltage to VDDC VL to VDDC Switch Resistance 6V < VVDDM < 11V, 0.1mA < ILOAD < 10mA 3.5V < VVDDM < 11V, VVDDC = 0 VL rising, 40mV hysteresis VL rising, 100mV hysteresis 2.25 2.3 2.40 2.4 2.83 3.00 3.12 3 2.50 2.5 7 V % V V VONM = 0 VFBM = 1.5V, VVDDC = 0 VFBM = 1.5V, VVDDC = 3V VFBM = 1.5V, VVDDC = 3V VFBM = VFBC = 1.5V, VONC = 3V VFBM = VFB1 = 1.5V, VON1 = 3V VFBM = VFB2 = 1.5V, VDCON2 = 3V VFBM = VFB3 = 1.5V, VDCON3 = 3V VFBM = VFBC = VFB1 = VFB2 = VFB3 = 1.5V, VONC = VON1 = VDCON2 = VDCON3 = 3V 3 370 35 270 410 470 470 470 960 20 600 55 450 700 750 750 750 1700 A A A A A A A A VIN 2.5 11 V SYMBOL CONDITIONS MIN TYP MAX UNITS
2
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Digital Camera Step-Down Power Supply
ELECTRICAL CHARACTERISTICS (continued)
(Circuit of Figure 1, VVDDM = 6V, VVDDC = 3V, PGNDM = PGND = GND, DCON1 = REF, VONM = 3V, VONC = VON1 = VDCON2 = VDCON3 = 0, TA = 0C to +85C, unless otherwise noted. Typical values are at TA = +25C.)
PARAMETER REFERENCE Reference Output Voltage REF Load Regulation REF Line Rejection REF Undervoltage Lockout Threshold OSCILLATOR OSC Discharge Trip Level OSC Input Bias Current OSC Discharge Resistance OSC Discharge Pulse Width LOGIC INPUTS (ONM, ONC, ON1) Input Low Level Input High Level Input Leakage Current MAIN DC-DC CONVERTER Main Output Voltage Adjust Range Main Idle ModeTM Threshold Main Current-Sense Amplifier Voltage Gain Main N Channel Turn-Off Threshold Main Slope Compensation Gain MAIN ERROR AMPLIFIER FBM Regulation Voltage FBM to COMPM Transconductance FBM Input Leakage Current COMPM Minimum Output Voltage COMPM Maximum Output Voltage VCOMPM(MAX) GEA Unity gain configuration, FBM = COMPM Unity gain configuration, FBM = COMPM, -5A < ILOAD < 5A VFBM = 1.35V VFBM = 1.35V, COMPM open VFBM = 1.15V, COMPM open 0.3 2.00 2.14 2.27 1.233 70 1.248 100 5 1.263 160 100 V S nA V V AVSWM AVCSM VOUT VOSC = 0.625V, measured between VDDM and LXM Measured between VDDM and LXM Measured between LXM and PGNDM 2.7 8 8.4 -26 0.16 20 9.3 -17 0.20 5.5 32 10.2 -8 0.24 V mV V/V mV V/V VIL VIH ONM ONC, ON1 ONM: VIN = 0 or 11V; ONC, ON1: VIN = 0 or 5V 1.8 1.6 0.01 1 0.4 V V A OSC rising VOSC = 1.1V VOSC = 1.5V 1.225 1.250 0.2 30 100 1.275 100 100 V nA ns VREF IREF = 20A 10A < IREF < 200A 2.7V < VOUT < 5.5V REF rising, 20mV hysteresis 0.9 1.235 1.248 5 1 1 1.260 9 5 1.1 V mV mV V SYMBOL CONDITIONS MIN TYP MAX UNITS
MAX1802
Idle Mode is a trademark of Maxim Integrated Products. _______________________________________________________________________________________ 3
Digital Camera Step-Down Power Supply MAX1802
ELECTRICAL CHARACTERISTICS (continued)
(Circuit of Figure 1, VVDDM = 6V, VVDDC = 3V, PGNDM = PGND = GND, DCON1 = REF, VONM = 3V, VONC = VON1 = VDCON2 = VDCON3 = 0, TA = 0C to +85C, unless otherwise noted. Typical values are at TA = +25C.)
PARAMETER MAIN SOFT-START Soft-Start Interval MAIN DRIVERS (DHM, DLM) Output Low Voltage Output High Voltage Driver Resistance Drive Current CORE DC-DC CONVERTER (VONC = 3V) Core Output Voltage Adjust Range Core Idle Mode Threshold Core Current-Sense Amplifier Transresistance Core Slope Compensation Gain FBC Regulation Voltage FBC to COMPC Transconductance FBC Input Leakage Current COMPC Minimum Output Voltage COMPC Maximum Output Voltage CORE SOFT-START (VONC = 3V) Soft-Start Interval CORE POWER SWITCHES (VONC = 3V) LXC Leakage Current Switch On-Resistance P-Channel Current Limit N-Channel Turn-Off Current RDSN RDSP VLXC = 0, 5.5V N-channel, ILXC = 0.75A P-channel, ILXC = 0.75A VOSC = 0.625V 18 0.01 150 180 0.75 100 180 20 350 400 A m A mA 1024 OSC cycles VCOMPM(MAX) GEA RCSC AVSWC Unity gain configuration, FBC = COMPC Unity gain configuration, FBC = COMPC, -5A < ILOAD < 5A VFBC = 1.35V VFBC = 1.35V, COMPC open VFBC = 1.15V, COMPC open 0.3 2.00 2.14 2.27 VOUT VOSC = 0.625V 1.25 70 0.7 0.16 1.233 70 190 1.0 0.20 1.248 100 5 5.5 320 1.3 0.24 1.263 160 100 V mA V/A V/V V S nA V V ISINK = 10mA ISOURCE = 10mA IDHM = 10mA, IDLM = 10mA Sourcing or sinking, VDHM or VVL = VVDDM / 2 VVDDM 0.11 4 400 11 0.11 V V mA OSC falling edge 1024 OSC cycles SYMBOL CONDITIONS MIN TYP MAX UNITS
CORE ERROR AMPLIFIER (VONC = 3V)
4
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Digital Camera Step-Down Power Supply
ELECTRICAL CHARACTERISTICS (continued)
(Circuit of Figure 1, VVDDM = 6V, VVDDC = 3V, PGNDM = PGND = GND, DCON1 = REF, VONM = 3V, VONC = VON1 = VDCON2 = VDCON3 = 0, TA = 0C to +85C, unless otherwise noted. Typical values are at TA = +25C.)
PARAMETER INTERNAL CLOCK OSC Clock Low Trip Level OSC Clock High Trip Level Maximum Duty Cycle Adjustment Range Maximum Duty Cycle Default Maximum Duty Cycle DCON_ Input Leakage Current DCON_ Input Sleep-Mode Threshold AUXILIARY ERROR AMPLIFIER FB_ Regulation Voltage FB_ to COMP_ Transconductance FB_ Input Leakage Current AUXILIARY DRIVERS (DL1, DL2, DL3) DL_ Driver Resistance DL_ Drive Current AUXILIARY SOFT-START Soft-Start Interval AUXILIARY SHORT-CIRCUIT PROTECTION Fault Interval 1024 OSC cycles 1024 OSC cycles Output high or low Sourcing or sinking, VDL_ = VVDDC / 2 4 400 11 mA GEA Unity gain configuration, FB_ = COMP_ Unity gain configuration, FB_ = COMP_, -5A < ILOAD < 5A VFB_ = 1.35V 1.233 70 1.248 100 5 1.263 160 100 V s nA VDCON _ = 0.625V VDCON _ = 1.25V to VVL VDCON _ = 0V to 3V VDCON _ rising, 50mV hysteresis 0.35 OSC falling edge VDCON _ = 0.625V VDCON _ = 1.25V to VVL 0.2 0.575 1.00 40 43 76 0.01 0.4 1 0.45 0.25 0.625 1.05 0.3 0.675 1.10 90 V V % % % A V SYMBOL CONDITIONS MIN TYP MAX UNITS
MAX1802
AUXILIARY DC-DC CONTROLLERS 1, 2, 3 (VON1 = VCON_ = 3V)
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Digital Camera Step-Down Power Supply MAX1802
ELECTRICAL CHARACTERISTICS
(Circuit of Figure 1, VVDDM = 6V, VVDDC = 3V, PGNDM = PGND = GND, DCON1 = REF, VONM = 3V, VONC = VON1 = VDCON2 = VDCON3 = 0, TA = -40C to +85C, unless otherwise noted.) (Note 1)
PARAMETER GENERAL Input Voltage Range SUPPLY CURRENT Shutdown Supply Current (from VDDM and VDDC) Main DC-DC Converter Supply Current (from VDDM) Main DC-DC Converter Supply Current (from VDDC) Main plus Core Supply Current (from VDDC) Main plus Auxiliary 1 Supply Current (from VDDC) Main plus Auxiliary 2 Supply Current (from VDDC) Main plus Auxiliary 3 Supply Current (from VDDC) Total Supply Current (from VDDC) VL REGULATOR VL Output Voltage VL Supply Rejection VL Undervoltage Lockout Threshold VL Switchover Voltage to VDDC VL to VDDC Switch Resistance REFERENCE Reference Output Voltage REF Load Regulation REF Line Rejection REF Undervoltage Lockout Threshold OSCILLATOR OSC Discharge Trip Level OSC Input Bias Current OSC Discharge Resistance OSC rising VOSC = 1.1V VOSC = 1.5V 1.225 1.275 100 100 V nA VREF IREF = 20A 10A < IREF < 200A 2.7V < VOUT < 5.5V REF rising, 20mV hysteresis 0.9 1.230 1.262 9 5 1.1 V mV mV V 6V < VVDDM < 11V, 0.1mA < ILOAD < 10mA 3.5V < VVDDM < 11V, VVDDC = 0 VL rising, 40mV hysteresis VL rising, 100mV hysteresis 2.25 2.3 2.83 3.12 3 2.50 2.5 7 V % V V VONM = 0 VFBM = 1.5V, VVDDC = 0 VFBM = 1.5V, VVDDC = 3V VFBM = 1.5V, VVDDC = 3V VFBM = VFBC = 1.5V, VONC = 3V VFBM = VFB1 = 1.5V, VON1 = VDCON1 = 3V VFBM = VFB2 = 1.5V, VDCON2 = 3V VFBM = VFB3 = 1.5V, VDCON3 = 3V VFBM = VFBC = VFB1 = VFB2 = VFB3 = 1.5V, VONC = VON1 = VDCON1 = VDCON2 = VDCON3 = 3V 20 600 55 450 700 750 750 750 A A A A A A A VIN 2.5 11 V SYMBOL CONDITIONS MIN TYP MAX UNITS
1700
A
6
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Digital Camera Step-Down Power Supply MAX1802
ELECTRICAL CHARACTERISTICS (continued)
(Circuit of Figure 1, VVDDM = 6V, VVDDC = 3V, PGNDM = PGND = GND, DCON1 = REF, VONM = 3V, VONC = VON1 = VDCON2 = VDCON3 = 0, TA = -40C to +85C, unless otherwise noted.) (Note 1)
PARAMETER LOGIC INPUTS (ONM, ONC, ON1) Input Low Level Input High Level Input Leakage Current MAIN DC-DC CONVERTER Main Output Voltage Adjust Range Main Idle Mode Threshold Main Current-Sense Amplifier Voltage Gain Main Zero-Crossing Threshold Main Slope Compensation Gain MAIN ERROR AMPLIFIER FBM Regulation Voltage FBM to COMPM Transconductance FBM Input Leakage Current COMPM Minimum Output Voltage COMPM Maximum Output Voltage MAIN DRIVERS (DHM, DLM) Output Low Voltage Output High Voltage Driver Resistance CORE DC-DC CONVERTER (VONC = 3V) Core Output Voltage Adjust Range Core Idle Mode Threshold Core Current-Sense Amplifier Transresistance Core Slope Compensation Gain FBC Regulation Voltage FBC to COMPC Transconductance GEA RCSC AVSWC Unity gain configuration, FBC = COMPC Unity gain configuration, FBC = COMPC, -5A < ILOAD < 5A VOUT VOSC = 0.625V 1.25 40 0.7 0.16 1.230 70 5.5 360 1.3 0.24 1.265 160 V mA V/A V/V V S ISINK = 10mA ISOURCE = 10mA IDHM = 10mA, IDLM = 10mA VVDDM 0.11 11 0.11 V V GEA Unity gain configuration, FBM = COMPM Unity gain configuration, FBM = COMPM, -5A < ILOAD < 5A VFBM = 1.35V VFBM = 1.35V, COMPM open VCOMPM(MAX) VFBM = 1.15V, COMPM open 0.3 2.00 2.27 1.230 70 1.265 160 100 V S nA V V AVSWM AVCSM VOUT VOSC = 0.625V, measured between VDDM and LXM Measured between VDDM and LXM Measured between LXM and PGNDM 2.7 2 8.4 -20 0.16 5.5 35 10.2 -8 0.24 V mV V/V mV V/V VIL VIH ONM ONC, ON1 ONM: VIN = 0 or 11V; ONC, ON1: VIN = 0 or 5V 1.8 1.6 1 0.4 V V A SYMBOL CONDITIONS MIN TYP MAX UNITS
CORE ERROR AMPLIFIER (VONC = 3V)
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Digital Camera Step-Down Power Supply MAX1802
ELECTRICAL CHARACTERISTICS (continued)
(Circuit of Figure 1, VVDDM = 6V, VVDDC = 3V, PGNDM = PGND = GND, DCON1 = REF, VONM = 3V, VONC = VON1 = VDCON2 = VDCON3 = 0, TA = -40C to +85C, unless otherwise noted.) (Note 1)
PARAMETER FBC Input Leakage Current COMPC Minimum Output Voltage COMPC Maximum Output Voltage LXC Leakage Current Switch On-Resistance N-Channel Turn-Off Current AUXILIARY DC-DC CONTROLLERS 1, 2, 3 (VON1 = VDCON_= 3V) INTERNAL CLOCK OSC Clock Low Trip Level OSC Clock High Trip Level Maximum Duty Cycle Adjustment Range DCON_ Input Leakage Current DCON_ Input Sleep-Mode Threshold AUXILIARY ERROR AMPLIFIER FB_ Regulation Voltage FB_ to COMP_ Transconductance FB_ Input Leakage Current AUXILIARY DRIVERS (DL1, DL2, DL3) DL_ Driver Resistance Output high or low 11 GEA Unity gain configuration, FB_ = COMP_ Unity gain configuration, FB_ = COMP_, -5A < ILOAD < 5A VFB_ = 1.35V 1.230 70 1.265 160 100 V s nA VDCON_ = 0V to 3V VDCON_ rising, 50mV hysteresis 0.35 OSC falling edge VDCON_ = 0.625V VDCON_ = 1.25V to VVL 0.2 0.575 1.00 40 0.3 0.675 1.10 90 1 0.45 % A V V V RDSN RDSP VCOMPC(MAX) SYMBOL VFBC = 1.35V VFBC = 1.35V, COMPC open VFBC = 1.15V, COMPC open 0.3 2.00 2.27 CONDITIONS MIN TYP MAX 100 UNITS nA V V
CORE POWER SWITCHES (VONC = 3V) VLXC = 0, 5.5V N-channel, ILXC = 0.75A P-channel, ILXC = 0.75A 5 20 350 400 190 A m mA
Note 1: Specifications to -40C are guaranteed by design and not production tested.
8
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Digital Camera Step-Down Power Supply
Typical Operating Characteristics
(Circuit of Figure 1, VVDDM = 6V, VVDDC = 3.3V, VONM = 3V, VONC = VON1 = VDCON2 = VDCON3 = 0, TA = +25C, unless otherwise noted.)
EFFICIENCY vs. LOAD CURRENT (MAIN CONVERTER)
MAX1802 toc01
MAX1802
EFFICIENCY vs. LOAD CURRENT (MAIN CONVERTER)
MAX1802 toc02
EFFICIENCY vs. LOAD CURRENT (CORE CONVERTER)
90 80 EFFICIENCY (%) 70 60 50 40 30 20 VIN = +2.5V VIN = +3.3V VIN = +5V
MAX1802 toc03
100 90 80 EFFICIENCY (%) 70 60 50 40 30 20 10 0 1 10 100 1000 VOUT = 3.3V VIN = +5V VIN = +7.2V VIN = +11V
100 90 80 EFFICIENCY (%) 70 60 50 40 30 20 10 0 VOUT = +5V 1 10 100 1000 VIN = +7.2V VIN = +11V
100
10 0 10,000 1
VOUT = +1.8V 10 100 LOAD CURRENT (mA) 1000
10,000
LOAD CURRENT (mA)
LOAD CURRENT (mA)
EFFICIENCY vs. LOAD CURRENT (CORE CONVERTER)
MAX1802 toc04
MAXIMUM DUTY CYCLE vs. VDCON_
MAX1802 toc05
DEFAULT MAXIMUM DUTY CYCLE vs. FREQUENCY
COSC = 470pF DEFAULT MAXIMUM DUTY CYCLE (%)
MAX1802 toc06
100 90 80 EFFICIENCY (%) 70 60 50 40 30 20 10 0 1 10 100 LOAD CURRENT (mA) VOUT = +2.5V VIN = +5V VIN = +3.3V
100
100
MAXIMUM DUTY CYCLE (%)
80
80
60
60
40
40
20
20
0 1000 0.4 0.5 0.6 0.7 0.8 0.9 1.0 1.1 1.2 VDCON_ (V)
0 0 200 400 600 800 1000 FREQUENCY (kHz)
OSCILLATOR FREQUENCY vs. ROSC
MAX1802 toc07
SHUTDOWN CURRENT vs. INPUT VOLTAGE
MAX1802 toc08
1000 OSCILLATOR FREQUENCY (kHz) COSC = 470pF COSC = 220pF 600 COSC = 100pF COSC = 47pF 400
10
SHUTDOWN CURRENT (A)
800
8
6
4
200
2
0 1 10 ROSC (k) 100 1000
0 0 2 4 6 8 10 12 INPUT VOLTAGE (V)
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Digital Camera Step-Down Power Supply MAX1802
Typical Operating Characteristics (continued)
(Circuit of Figure 1, VVDDM = 6V, VVDDC = 3.3V, VONM = 3V, VONC = VON1 = VDCON2 = VDCON3 = 0, TA = +25C, unless otherwise noted.)
REFERENCE VOLTAGE vs. TEMPERATURE
MAX1802 toc09
REFERENCE VOLTAGE vs. REFERENCE CURRENT
MAX1802 toc10
1.260
1.253 1.252 REFERENCE VOLTAGE (V) 1.251 1.250 1.249 1.248
REFERENCE VOLTAGE (V)
1.255
1.250
1.245
1.240 -40
1.247 -20 0 20 40 60 80 0 50 100 150 200 250 TEMPERATURE (C) REFERENCE CURRENT (A)
FB_ TO COMP_ SMALL-SIGNAL OPEN-LOOP FREQUENCY RESPONSE
MAX1802 toc11
MAIN OUTPUT STARTUP RESPONSE
MAX1802 toc12
60 SMALL-SIGNAL RESPONSE (dB) 50 40 30 20 10 0 1 10 100 FREQUENCY (kHz) 1000
VONM 5V/div 0V VMAIN 2V/div IOUT 200mA/div
0V
0A
10,000
1ms/div
CORE OUTPUT STARTUP RESPONSE
MAX1802 toc13
AUXILIARY CONTROLLER STARTUP RESPONSE
MAX1802 toc14
VONC 5V/div 0V VCORE 2V/div 0V IOUT 100mA/div
0V
VON_ 5V/div VOUT 2V/div
0V
IOUT 200mA/div 0A 0A 1ms/div 1ms/div
10
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Digital Camera Step-Down Power Supply
Typical Operating Characteristics (continued)
(Circuit of Figure 1, VVDDM = 6V, VVDDC = 3.3V, VONM = 3V, VONC = VON1 = VDCON2 = VDCON3 = 0, TA = +25C, unless otherwise noted.)
STARTUP SEQUENCE
MAX1802 toc15
MAX1802
MAIN OUTPUT LOAD-TRANSIENT RESPONSE
MAX1802 toc16
VONM 5V/div 0V VMAIN 2V/div 0V VCORE 2V/div
VOUT AC-COUPLED 100mV/div
ILOAD 200mA/div
0A 1ms/div
0A COUT = 100F 400s/div
CORE OUTPUT LOAD-TRANSIENT RESPONSE
MAX1802 toc17
AUXILIARY OUTPUT LOAD-TRANSIENT RESPONSE
MAX1802 toc18
VOUT AC-COUPLED 200mV/div
VOUT AC-COUPLED 100mV/div
ILOAD 100mA/div ILOAD 200mA/div 0A VOUT = 2.5V 500s/div 400s/div 0A
MAIN TRANSIENT RESPONSE SUBJECT TO CORE TRANSIENT
MAX1802 toc19
VOUT (MAIN) AC-COUPLED 20mV/div
ILOAD (CORE) 100mA/div 0A VOUT = 2.5V 2.5ms/div
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11
Digital Camera Step-Down Power Supply MAX1802
Pin Description
PIN 1 2 NAME FBM COMPM FUNCTION Main DC-DC Converter Feedback Input. Connect a feedback resistive voltage-divider from the output to FBM to set the main output voltage. Regulation voltage is VREF (1.25V). Compensation for Main Controller. Output of main transconductance error amplifier. Connect a series resistor and capacitor to GND to compensate the main control loop (see Compensation Design). Main Converter Enable Input. High level turns on the main converter and VL regulator. Connect ONM to VDDM to automatically start the converter. When the main converter is off, all other outputs are disabled. Internal Bias Voltage. VH provides bias to the main controller. Bypass VH to VDDM with a 0.1F or greater ceramic capacitor. Battery Input. VDDM supplies power to the IC and also serves as a high-side current-sense input for the main DC-DC controller. Connect VDDM as close as possible to the source of the external P-channel switching MOSFET for the main controller. External P-Channel MOSFET Gate-Drive Output for Main Controller. DHM swings between VDDM and PGNDM with 400mA (typ) drive current. Connect DHM to the gate of the external P-channel switching MOSFET for the main controller. Main DC-DC Controller Current-Sense Input. Connect LXM to the drains of the external P- and Nchannel switching MOSFETs for the main converter. LXM serves as the current-sense input for both P- and N-channel switching MOSFETs. Connect LXM as close as possible to the drain of the external P-channel switching MOSFET for the main controller. External N-Channel MOSFET Gate-Drive Output for Main Controller. DLM swings between VDDM and PGNDM with 400mA (typ) drive current. Connect DLM to the gate of the external N-channel switching MOSFET for the main controller. Power Ground for Main DC-DC Controller. PGNDM also serves as a low-side current-sense input for the main DC-DC controller. Connect PGNDM as close as possible to the source of the external N-channel switching MOSFET for the main controller. Oscillator Control. Connect a timing capacitor from OSC to GND and a timing resistor from OSC to VL to set the switching frequency between 100kHz and 1MHz (see Setting the Switching Frequency). Maximum Duty Cycle Control Input for Auxiliary Controller 1. Connect DCON1 to VL to set the default maximum duty cycle. Connect a resistive voltage-divider from REF to DCON1 to set the maximum duty cycle between 40% and 90%. Pull DCON1 below 300mV to turn the controller off. External MOSFET Gate Drive Output for Auxiliary Controller 1. DL1 swings between VDDC and PGND with 400mA (typ) drive current. Connect DL1 to the gate of the external switching N-channel MOSFET for auxiliary controller 1. Enable Input for Auxiliary Controller 1. Connect ON1 to VL to automatically start auxiliary controller 1. Compensation for Auxiliary Controller 1. Output of auxiliary controller 1 transconductance error amplifier. Connect a series resistor and capacitor from COMP1 to GND to compensate the auxiliary controller 1 control loop (see Compensation Design). Feedback Input for Auxiliary Controller 1. Connect a feedback resistive voltage-divider from the output of auxiliary controller 1 to FB1 to set the output voltage. Regulation voltage is VREF (1.25V). Feedback Input for Auxiliary Controller 2. Connect a feedback resistive voltage-divider from the output of auxiliary controller 2 to FB2 to set the output voltage. Regulation voltage is VREF (1.25V).
3
ONM
4
VH
5
VDDM
6
DHM
7
LXM
8
DLM
9
PGNDM
10
OSC
11
DCON1
12 13 14
DL1 ON1 COMP1
15 16
FB1 FB2
12
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Digital Camera Step-Down Power Supply
Pin Description (continued)
PIN NAME FUNCTION Compensation for Auxiliary Controller 2. Output of auxiliary controller 2 transconductance error amplifier. Connect a series resistor and capacitor from COMP2 to GND to compensate the auxiliary controller 2 control loop (see Compensation Design). Maximum Duty Cycle Control Input for Auxiliary Controller 2. Connect DCON2 to VL to set the default maximum duty cycle. Connect a resistive voltage-divider from REF to DCON2 to set the maximum duty cycle between 40% and 90%. Pull DCON2 below 300mV to turn the controller off. External MOSFET Gate Drive Output for Auxiliary Controller 2. DL2 swings between VDDC and PGND with 400mA (typ) drive current. Connect DL2 to the gate of the external switching N-channel MOSFET for auxiliary controller 2. External MOSFET Gate Drive Output for Auxiliary Controller 3. DL3 swings between VDDC and PGND with 400mA (typ) drive current. Connect DL3 to the gate of the external switching N-channel MOSFET for auxiliary controller 3. Compensation for Auxiliary Controller 3. Output of auxiliary controller 3 transconductance error amplifier. Connect a series resistor and capacitor from COMP3 to GND to compensate the auxiliary controller 3 control loop (see Compensation Design). Feedback Input for Auxiliary Controller 3. Connect a feedback resistive voltage-divider from the output of auxiliary controller 3 to FB3 to set the output voltage. Regulation voltage is VREF (1.25V). Maximum Duty Cycle Control Input for Auxiliary Controller 3. Connect DCON3 to VL to set the default maximum duty cycle. Connect a resistive voltage-divider from REF to DCON3 to set the maximum duty cycle between 40% and 90%. Pull DCON3 below 300mV to turn the controller off. Core Converter Enable Input. High level turns on the core converter. Connect ONC to VL to automatically start the core converter. Power Ground. Sources of internal N-channel MOSFET power switches. Connect PGND to GND as close to the IC as possible. Core Power Switching Node. Drains of the internal P- and N-channel MOSFET switches for the core converter. Core DC-DC Converter Power Input. VDDC is connected to the source of the internal P-channel MOSFET power switch for the core converter. VDDC is limited to 5.5V. For battery voltages greater than 5.5V, connect VDDC to the main output. Bypass VDDC to PGND with a 1F or greater ceramic capacitor. Internal Low-Voltage Bypass. The internal circuitry is powered from VL. An internal linear regulator powers VL from VDDM when VDDC is less than 2.4V. When VDDC is greater than 2.4V, an internal switch connects VL to VDDC. Bypass VL to GND with a 1.0F or greater ceramic capacitor. Compensation for Core Converter. Output of core transconductance error amplifier. Connect a series resistor and capacitor to GND to compensate the core control loop (see Compensation Design). Core DC-DC Converter Feedback Input. Connect a feedback resistive voltage-divider from the core output to FBC to set the output voltage. Regulation voltage is VREF (1.25V). 1.25V Reference Output. Bypass REF to GND with a 0.1F or greater ceramic capacitor. Analog Ground
MAX1802
17
COMP2
18
DCON2
19
DL2
20
DL3
21
COMP3
22
FB3
23
DCON3
24 25 26
ONC PGND LXC
27
VDDC
28
VL
29 30 31 32
COMPC FBC REF GND
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Digital Camera Step-Down Power Supply MAX1802
Detailed Description
The MAX1802 typical application circuit is shown in Figure 1. It features two step-down DC-DC converters (main and core), three auxiliary step-up DC-DC controllers, and control capability for multiple external MAX1801 slave DC-DC controllers. Together, these provide a complete high-efficiency power-supply solution for digital still cameras. Figures 2 and 3 show the MAX1802 functional block diagrams. drain of the external P-channel MOSFET. The MAX1802 main output switches to Idle Mode at light loads to improve efficiency by leaving the P-channel switch on until the voltage across the MOSFET reaches the 20mV Idle Mode threshold. The Idle Mode current is 20mV divided by the MOSFET on-resistance. By forcing the inductor current above the Idle Mode threshold, more energy is supplied to the output capacitor than is required by the load. The switch and synchronous rectifiers then remain off until the output capacitor discharges to the regulation voltage. This causes the converter to operate at a lower effective switching frequency at light loads, thus improving efficiency. An internal comparator turns off the N-channel synchronous rectifier as the inductor current drops near zero, by measuring the voltage across the MOSFET. If the Nchannel MOSFET on-resistance is low (less than that of the P-channel switch), it may cause the MOSFET to turn off prematurely, degrading efficiency. This is especially critical for high input voltage applications, such as with 2 series Li+ cells. In this case, use an N-channel MOSFET with greater on-resistance than the P-channel switch, and/or place a Schottky recitifier across the Nchannel MOSFET gate-source. The voltage at COMPM is typically clamped to VCOMPM(MAX) = 2.14V, thereby limiting the inductor current. The peak inductor current (ILIM) and the maximum average output current (I OUT(MAX)) are determined by the following equations: V A VCOMPM(MAX) - VREF 1 + OUT VSWM VIN ILIM = A VCSM RDSP 1 - IOUT(MAX) = ILIM - VOUT VOUT VIN 2 fOSC L
Master-Slave Configuration
The MAX1802 supports MAX1801 "slave" controllers that obtain input power, a voltage reference, and an oscillator signal directly from the MAX1802 "master" DC-DC converter. The master-slave configuration reduces system cost by eliminating redundant circuitry and controlling the harmonic content of noise with synchronized converter switching.
Main DC-DC Converter
The MAX1802 main step-down DC-DC converter generates a 2.7V to 5.5V output voltage from a 2.5V to 11V battery input voltage. When the battery voltage is lower than the main regulation voltage, the regulator goes into dropout and the P-channel switch remains on. In this condition, the output voltage is slightly lower than the input voltage. The converter drives an external Pchannel MOSFET power switch and an external Nchannel MOSFET synchronous rectifier. The converter operates in a low-noise, constant-frequency PWM current mode to regulate the voltage across the load. Switching harmonics generated by fixed-frequency operation are consistent and easily filtered. The external P-channel MOSFET switch turns on during the first part of each cycle, allowing current to ramp up in the inductor and store energy in a magnetic field while supplying current to the load. During the second part of each cycle, the P-channel MOSFET turns off and the voltage across the inductor reverses, forcing current through the external N-channel synchronous rectifier to the output filter capacitor and load. As the energy stored in the inductor is depleted, the current ramps down. The synchronous rectifier turns off when the inductor current approaches zero or at the beginning of a new cycle, at which time the P-channel switch turns on again. The current-mode PWM converter uses the voltage at COMPM to program the inductor current and regulate the output voltage. The converter detects inductor current by sensing the voltage across the source and
where AVSWM is the main slope compensation gain (0.20V/V), AVCSM is the voltage gain of the main current-sense amplifier (9.3V/V), RDSP is the on-resistance of the external P-channel MOSFET switch, and L is the inductor value. Note that the current limit increases as the input/output voltage ratio increases.
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INPUT 2.5V TO 11V VL 40.2k 1F 1.1M CCD BIAS 1F 10 OSC DL1 D2 Q2 12 COSC 31 100pF REF FB1 0.1F 11 DCON1 D3 +18V 1F 19 Q3 FB2 DHM 24 ONC ON1 DLM 3 OFF 2 29 RCM 33k RCC 90k CCC 470pF CC1 1000pF RC1 10k RC2 10k CC2 1000pF RC3 10k CC3 1000pF GND 32 PGND 25 CCM 4.7nF 14 ONM COMPM FBM COMPC VDDC COMP1 17 COMP2 21 COMP3 FBC 30 LXC 26 44.2k PGNDM 13 LXM 16 100k 6 7 8 9 1 27 10H 10F 1F +1.8V CORE Q4 10H Q5 D6 100F 165k 100k +3.3V MAIN D4 LCD BIAS +12V 1F 1.34M DCON2 DCON3 DL2 Q1 20 DL3 22 FB3 MAX1802 VL 100k 1F 28 18 D5 +7V BACKLIGHT 4.7F 464k 23 15 100k -7.5V 0.1H 4.7H ROSC 7 8 DL
IN OSC
10F VL 5 VDDM VH 4 0.1F +15V
D1
+5V MOTOR DRIVE 1
Figure 1. Typical Application Circuit
MAX1801
6
3 REF COMP 4 GND DCON 5 2
ON
MAX1802
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100k
Q1, Q2, Q3: FDN337N Q4, Q5: SEE MOSFET SELECTION SECTION D1, D2, D3, D4: CMSD-4448 D5: MBR0502L
Digital Camera Step-Down Power Supply
15
Digital Camera Step-Down Power Supply MAX1802
OSC CLOCK GENERATOR CLK COMPM VREF 100ns ONE-SHOT REFERENCE REF
CLK
VH
VH VDDM DHM
FBM
MAIN CURRENT-MODE DC-DC CONTROLLER
LXM DLM
SOFT-START ONM
VREF VL LDO 2.4V CLK
PGNDM VL GND
COMPC FBC CORE CURRENT MODE DC-DC CONTROLLER VREF ONC SOFT-START
VDDC
LXC
PGND
Figure 2. Simplified Block Diagram, Including Main and Core
Core DC-DC Converter
The MAX1802 core step-down DC-DC converter generates a 1.25V to 5.5V output voltage from the main controller output. The core converter has the same low-noise, constant-frequency PWM current-mode architecture as the main controller. However, it uses an internal P-channel MOSFET power switch and N-channel MOSFET synchronous rectifier to maximize efficiency and reduce circuit size and external component count. The core converter internally monitors the inductor current for current-mode regulation of the output voltage, as well as overload protection, automatic Idle Mode switchover, and turning off the synchronous rectifier when the inductor current approaches zero. By switching to Idle Mode at light loads and turning the synchronous rectifier off at zero current, light-load efficiency is improved. The core converter is inactive until the main output has started. The voltage at COMPC is typically clamped to VCOMPC(MAX) = 2.14V, thereby limiting the inductor current. The peak inductor current limit (ILIM) and the maximum average output current (I OUT(MAX) ) are determined by the following equations:
16
V A VCOMPC(MAX) - VREF 1 + OUT VSWC VIN ILIM = RCSC V 1 - OUT VOUT VIN IOUT(MAX) = ILIM - 2 fOSC L where A VSWC is the core slope compensation gain (0.20V/V), RCSC is the transresistance of the core current-sense amplifier (1V/A), and L is the inductor value. Note that the current limit increases as the input/output ratio increases.
Auxiliary DC-DC Controllers
The MAX1802's three auxiliary controllers operate in a low-noise, fixed-frequency, PWM mode with output power limited by the external components. The con-
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Digital Camera Step-Down Power Supply MAX1802
FB_
COMP_ R LEVEL SHIFT SOFTSTART Q DL_
S
REF DCON_
CLK OSC
FAULT PROTECTION
Figure 3. Auxiliary Controller Block Diagram
trollers regulate their output voltages by modulating the pulse width of the drive signal for an external N-channel MOSFET switch. The auxiliary controllers are inactive until the main output has started. Figure 3 shows a block diagram for a MAX1802 auxiliary PWM controller. The sawtooth oscillator signal at OSC governs the internal timing. At the beginning of each cycle, DL_ goes high to turn on the external MOSFET switch. The MOSFET switch turns off when the internally level-shifted sawtooth rises above COMP_ or when the maximum duty cycle is exceeded. The switch remains off until the beginning of the next cycle. An internal transconductance amplifier establishes an integrated error voltage at COMP_, thereby increasing the loop gain for improved regulation accuracy.
begins soft-start operation. The main DC-DC output reaches full regulation voltage after 1024 soft-start oscillator cycles. Once the main DC-DC converter completes soft-start, the core DC-DC converter and the auxiliary DC-DC controllers are enabled. As the voltage at VDDC rises above 2.4V, the internal linear regulator turns off and an internal 3 switch connects VL directly to VDDC, which is typically connected to the output of the main DC-DC converter. The core DC-DC converter and the auxiliary DC-DC controllers have independent on-off control and softstart. The main DC-DC converter shuts down with a low input at ONM. The core DC-DC converter shuts down with a low input at ONC. Turn auxiliary DC-DC converter 1 off by driving either ON1 or DCON1 to GND. Turn off auxiliary controller 2 or 3 by driving DCON2 or DCON3 to GND.
Power-Up Sequence
The MAX1802 is in the shutdown state with all circuitry off when the ONM input is low (<1.3V). When ONM goes high, an internal linear regulator generates 3V at the VL output from the VDDM input to power internal circuitry. As VL rises above the 2.4V undervoltage lockout threshold, the internal reference and oscillator begin to function and the main DC-DC converter
Reference
The MAX1802 has an internal 1.248V, 1% reference. Connect a 0.1F bypass capacitor from REF to GND within 0.2in (5mm) of the REF pin. REF can source up to 200A of external load current, and it is enabled whenever ONM is high and VL is above the undervolt17
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Digital Camera Step-Down Power Supply MAX1802
age lockout threshold. The internal core converter, auxiliary controllers, and MAX1801 slave controllers each sink up to 30A REF current during startup. If multiple MAX1801 controllers are turned on simultaneously, ensure that the master voltage reference can provide sufficient current, or buffer the reference with an appropriate unity-gain amplifier.
VL
ROSC OSC COSC VREF (1.25V)
Oscillator
The oscillator uses a comparator, a 100ns one-shot, and an internal N-channel MOSFET switch in conjunction with an external timing resistor and capacitor to generate the oscillator signal at OSC (Figure 4). The capacitor voltage exponentially approaches VL from zero with a time constant given by the R OSC C OSC product when the switch is open, and the comparator output becomes high when the capacitor voltage reaches VREF (1.25V). At that time, the one-shot activates the internal MOSFET switch to discharge the capacitor within a 100ns interval, and the cycle repeats. Note that the oscillation frequency changes as VL changes during startup. The oscillation frequency is constant while the VL voltage is constant.
100ns ONE-SHOT
MAX1802
Maximum Duty Cycle
The MAX1802's three auxiliary controllers use the sawtooth oscillator signal generated at OSC, the voltage at DCON_, and an internal comparator to limit their maximum duty cycles (see Setting the Maximum Duty Cycle). Limiting the duty cycle can prevent saturation in some magnetic components. A low maximum duty cycle can also force the converter to operate in discontinuous current mode, simplifying design stability at the cost of a slight reduction in efficiency.
Figure 4. Oscillator
cation, short-circuit current is not limited, due to the DC current path through the inductor and output rectifier to the short circuit. If short-circuit protection is required in a step-up configuration, use a protection device such as a fuse to limit short-circuit current.
Design Procedure
Setting the Switching Frequency
Choose a switching frequency to optimize external component size or circuit efficiency for the particular MAX1802 application. Switching frequencies between 400kHz and 500kHz offer a good balance between component size and circuit efficiency. Higher frequencies allow smaller components, and lower frequencies improve efficiency. The switching frequency is set with an external timing resistor (ROSC) and capacitor (COSC). At the beginning of a cycle, the timing capacitor charges through the resistor until it reaches VREF. The charge time t1 is: t1 = -ROSC(COSC+10pF) In [1 - (VREF / VVL)] Once the voltage at OSC reaches VREF, it discharges through an internal switch over time t2 = 200ns. The oscillator frequency is fOSC = 1 / (t1 + t2). Set fOSC in the range 100kHz f OSC 1MHz. Choose C OSC between 47pF and 470pF. Determine ROSC from the relation:
Soft-Start
All the MAX1802 converters feature a soft-start function that limits inrush current and prevents excessive battery loading at startup by ramping the output voltage to the regulation voltage. This is achieved by increasing the internal reference inputs to the controller transconductance amplifiers from 0 to the 1.25V reference voltage over 1024 oscillator cycles when initial power is applied or when the controller is enabled.
Overload Protection
The MAX1802's three auxiliary controllers have fault protection that prevents damage to transformer-coupled or SEPIC circuits due to an output overload condition. When the output voltage drops out of regulation for 1024 oscillator clock periods, the auxiliary controller is disabled to prevent excessive output current. Restart the controller by cycling the voltage at ON_ or DCON_ to GND and back to the on state. For a step-up appli18
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Digital Camera Step-Down Power Supply
ROSC = (200ns - 1/fOSC) / (COSC + 10pF) ln (1 - VREF / VVL) See the Typical Operating Characteristics for fOSC vs. ROSC using different values of COSC. Due to duty cycle limitation in the main controller, keep fOSC VMAIN / (VVDDM(MAX) 500ns).
VOSC (V) 1.25 VDCON_ DMAX = tH tL + tH
MAX1802
Setting the Output Voltages
Set the MAX1802 output voltage of each converter by connecting a resistive voltage-divider from the output voltage to the corresponding FB_ input. The FB_ input bias current is <100nA, so choose RL (the low-side FB_-to-GND resistor) to be 100k. Choose R H (the high-side output-to-FB_ resistor) according to the relation: V RH = RL OUT - 1 1.248
0.25 0
CLK
tL
tH
Setting the Maximum Duty Cycle
The oscillator signal at OSC and the voltage at DCON_ are used to generate the internal clock signals for the three MAX1802 auxiliary controllers (CLK in Figure 3). The internal clock's falling edge occurs when VOSC exceeds VDCON_ (set by a resistive divider). The internal clock's rising edge occurs when VOSC falls below 0.25V (Figure 5). The adjustable maximum duty cycle range is 40% to 90% (see Maximum Duty Cycle vs. V DCON _ in the Typical Operating Characteristics). The maximum duty cycle defaults to 76% at 100kHz if VDCON_ is at or above the voltage at V REF (1.25V) (see Default Maximum Duty Cycle vs. Frequency in the Typical Operating Characteristics). The controller shuts down if VDCON_ is <0.3V.
Figure 5. Auxiliary Controller Internal Clock Signal Generation
Inductor Selection
Main and Core Step-Down Converters MAX1802 main and core step-down converters offer best efficiency when the inductor current is continuous. For most designs, a reasonable inductor value (LIDEAL) can be derived from the following equation, which sets continuous peak-to-peak inductor current at 1/3 the DC inductor current: 3 (VIN - VDSP ) D (1- D) LIDEAL = IOUT fOSC where D, the duty cycle, is given by: D= VOUT + VDSN VIN - VDSP + VDSN
In these equations, VDSP is the voltage drop across the P-channel MOSFET switch, and VDSN is the voltage drop across the N-channel MOSFET synchronous rectifier. Given LIDEAL, the consistent peak-to-peak inductor current is 0.33 IOUT. The maximum inductor current is 1.17 IOUT. Inductance values smaller than LIDEAL can be used; however, the maximum inductor current will rise as L is reduced, and a larger output capacitance will be required to maintain the same output ripple. For stable operation, the minimum inductance is limited by the internal slope compensation. The minimum inductor values for main and core are given by: 0.5 VOUT RDSP LMIN(MAIN) = 1 - DMAX 0.013 fOSC and 0.5 VOUT LMIN(CORE) = 1 - DMAX 0.13 fOSC where RDSP is the on-resistance of the P-channel MOSFET switch, and DMAX = VOUT / VIN. Auxiliary Step-Up Controllers The three MAX1802 auxiliary step-up controllers offer best efficiency when the inductor current is continuous.
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Digital Camera Step-Down Power Supply MAX1802
Use discontinuous current when the step-up ratio (VOUT / VIN) is greater than 1 / (1 - DMAX). Continuous Inductor Current A reasonable inductor value (LIDEAL) can be derived from the following equation, which sets continuous peak-to-peak inductor current at 1/3 the DC inductor current: LIDEAL = 3 VIN(MAX ) - VDSN D(1- D) IOUT fOSC The output capacitor is required to keep the output voltage ripple small and to ensure regulation control-loop stability. The output capacitor must have low impedance at the switching frequency. Tantalum and ceramic capacitors are good choices. Tantalum capacitors typically have high capacitance and medium-to-low equivalent series resistance (ESR) so that ESR dominates the impedance at the switching frequency. In turn, the output ripple is approximately: VRIPPLE IL(p-p) ESR where IL(p-p) is the peak-to-peak inductor current. Ceramic capacitors typically have lower ESR than tantalum capacitors, but with relatively small capacitance that dominates the impedance at the switching frequency. In turn, the output ripple is approximately: VRIPPLE IL(p-p) ZC where IL(p-p) is the peak-to-peak inductor current, and ZC 1 / (2 fOSC COUT ). See the Compensation Design section for a discussion of the influence of output capacitance and ESR on regulation control-loop stability. The capacitor voltage rating must exceed the maximum applied capacitor voltage. For most tantalum capacitors, manufacturers suggest derating the capacitor by applying no more than 70% of the rated voltage to the capacitor. Ceramic capacitors are typically used up to the voltage rating of the capacitor. Consult the manufacturer's specifications for proper capacitor derating.
(
)
where D, the duty cycle, is given by: D 1- VIN VOUT + VD
In these equations, VDSN is the voltage drop across the N-channel MOSFET switch, and VD is the forward voltage drop across the rectifier. Given LIDEAL, the consistent peak-to-peak inductor current is 0.33 IOUT / (1 - D). The maximum inductor current is 1.17 IOUT / (1 - D). Inductance values smaller than LIDEAL can be used; however, the maximum inductor current will rise as L is reduced, and a larger output capacitance will be required to maintain the same output ripple. The inductor current will become discontinuous if IOUT decreases by more than a factor of six from the value used to determine LIDEAL. Discontinuous Inductor Current In the discontinuous mode, each MAX1802 auxiliary controller regulates the output voltage by adjusting the duty cycle to allow adequate power transfer to the load. To ensure regulation under worst-case load conditions (maximum IOUT), choose: L= VOUT DMAX 2 IOUT fOSC
MOSFET Selection
The MAX1802 main converter and auxiliary controllers drive external logic-level P- and/or N-channel MOSFETs as the circuit switching elements. The key selection parameters are: * On-resistance (RDS(ON)) * Maximum drain-to-source voltage (VDS(MAX)) * Total gate charge (Qg) * Reverse transfer capacitance (CRSS) Because the main converter's external MOSFETs are used for current sense, they directly determine the output current capability and efficiency of the main converter. It is important to select the appropriate external MOSFETs for the main converter. The P-channel onresistance (RDSP) at minimum input voltage (VVDDM) must be low enough so that the converter can produce the desired output current as determined by the IOUT(MAX) equation in the Main DC-DC Converter section. The N-channel on-resistance (RDSN) determines
The peak inductor current is VIN DMAX / (L fOSC). The inductor's saturation current rating should meet or exceed the calculated peak inductor current.
Input and Output Filter Capacitors
The input capacitor (CIN) reduces the current peaks drawn from the battery or input power source. The impedance of the input capacitor at the switching frequency should be less than that of the input source so that high-frequency switching currents do not pass through the input source.
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Digital Camera Step-Down Power Supply
the N-channel turn-off current (equal to 17mV/RDSN). Choose RDSN value between RDSP and 3RDSP to keep the N-channel turn-off current low for optimal efficiency. If a lower RDSN is used, connect a Schottky diode from PGNDM to LXM for better efficiency (see Diode Selection). For the main converter, the external gate drive swings between the voltage at VDDM and GND. For the auxiliary controllers, the external gate drive swings between the voltage at VDDC and GND. Use a MOSFET whose on-resistance is specified at or below the minimum gate drive voltage swing, and make sure that the maximum voltage swing does not exceed the maximum gate-source voltage specification of the MOSFET. The gate charge, Qg, includes all capacitance associated with gate charging and helps to predict the transition time required to drive the MOSFET between on and off states. The power dissipated in the MOSFET is due to RDS(ON) and transition losses. The RDS(ON) loss is: P1 D IL2 RDS(ON) where D is the duty cycle, IL is the average inductor current, and RDS(ON) is the on-resistance of the MOSFET. The transition loss is approximately: P2 VSWING IL fOSC t T 3 high-temperature applications, use ultra-fast junction rectifiers.
MAX1802
Compensation Design
Each DC-DC converter has an internal transconductance error amplifier whose output is used to compensate the control loop. Typically, a series resistor and capacitor are inserted from COMP_ to GND to form a pole-zero pair. The external inductor, the output capacitor, the compensation resistor and capacitor, and for the main converter, the external P-channel MOSFET, govern control-loop stability. The inductor and output capacitor are usually chosen in consideration of performance, size, and cost, but the compensation resistor and capacitor are chosen to optimize control-loop stability. The component values in the circuit of Figure 1 yield stable operation over a broad range of input/output voltages and converter switching frequencies. Follow the procedures below for optimal compensation. In the following descriptions, Bode plots are used to graphically describe the loop response of the converters over frequency. The Bode plot shows loop gain and phase vs. frequency. A single pole results in a -20dB per decade slope and a -90 phase shift, and a single zero results in a +20dB per decade slope and a +90 phase shift. The stability of the system can be determined by the phase margin (how far from 0 the loop phase is when the response drops to 0dB) and gain margin (how far below 0dB the gain is when the phase reaches 0). The system is stable for phase margins >30, and a phase margin of 45 is preferred. The gain margin should be at least 10dB. Main Converter The main converter uses current mode to regulate the output voltage by forcing the required current through the inductor. Since the P-channel MOSFET operates with constant drain-source on-resistance (RDSP), the voltage across the MOSFET is proportional to the inductor current. The converter current-sense amplifier measures the "on" MOSFET drain-source voltage to determine the inductor current for regulation. The gain through the current-sense amplifier (measured across the MOSFET) is AVCSM = 9.3V/V. The voltage-divider attenuates the loop gain by AVDV = VREF / VOUT, and the gain DC voltage of the error amplifier is AVEA = 2000V/V. The controller forces the peak inductor current (IL) such that: IL RDSP AVCSM = VOUT AVDV AVEA or IL = VOUT AVDV AVEA / (AVCSM RDSP)
where VSWING is VOUT for the auxiliary controllers or VIN(MAX) for the main and core converters, IL is the average inductor current, fOSC is the converter switching frequency, and tT is the transition time. The transition time is approximately Qg / IG, where Qg is the total gate charge, and IG is the gate drive current (0.4A typ). The total power dissipation in the MOSFET is PMOSFET = P1 + P2.
Diode Selection
The main and core converters use synchronous rectifiers and thus do not require a diode. However, if the external N-channel synchronous rectifier has low onresistance (less than the P-channel on-resistance), the high N-channel turn-off current results in lower efficiency. In that case, connect a Schottky diode, rated for maximum output current, from PGNDM to LXM to improve efficiency. The auxiliary controllers require external rectifiers. For low-output-voltage applications, use a Schottky diode to rectify the output voltage because of the diode's low forward voltage and fast recovery time. Schottky diodes exhibit significant leakage current at high reverse voltages and high temperatures. Thus, for high-voltage,
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Digital Camera Step-Down Power Supply MAX1802
and the output voltage is IOUT RLOAD, which is equal to IL RLOAD. Thus, the total DC loop gain is: AVDC = RLOAD AVDV AVEA / (AVCSM RDSP) or AVDC = 215 VREF RLOAD / (VOUT RDS(ON)) Because of the current-mode control, there is a single pole in the loop response due to the output capacitor. This pole is at the frequency (in Hz): PO = 1 / (2 RLOAD COUT) Note that as the load resistance increases, the pole moves to a lower frequency. However, the DC loop gain increases by the same amount since they are both dependent on RLOAD. Thus, the crossover frequency (frequency at which the loop gain drops to 0dB), which is the product of the pole and the gain, remains at the same frequency. The compensation network creates a pole and zero at the frequencies (in Hz): PC = GEA / (4000 CC) = 1 / (4x107 CC) and ZC = 1 / (2 RC CC) and the ESR of the output filter capacitor causes a zero in the loop response at the frequency (in Hz): ZO = 1 / (2 COUT ESR) The DC gain and the poles and zeros are shown in the Bode plot of Figure 6. To achieve a stable circuit with the Bode plot of Figure 6, use the following procedure: 1) Determine the desired crossover frequency, either 1/3 of the zero due to the output capacitor ESR: fC = Z O / 3 = 1 6 COUT ESR
O Z0 FREQUENCY AVDC PC PHASE 90 GAIN (dB) ZC = PO PHASE MARGIN GAIN 0 PHASE 180
Figure 6. Current-Mode Step-Down Converter Bode Plot
or PO = ILOAD(MAX) 2 VOUT COUT
3) Determine the compensation resistor required to set the desired crossover frequency: RC = 20M fC A VDC PO
or, by simplifying and using the typical VREF = 1.25V: RC = 468k/V VOUT COUT RDSP fC 4) Determine the compensation capacitor to set the proper error-amplifier pole and zero determined from the above equations: CC = 1 2 RC PO
or 1/5 of the switching frequency: f fC = SW 5 whichever is lower. 2) Determine the pole frequency due to the output capacitor and the load resistor: PO = 1 2 RLOAD(MIN) COUT
Core Converter Compensating the core converter is similar to the compensation of the main converter described above. The only difference is that the current is measured internally, and the gain (transresistance) of the current-sense amplifier is RCSC = 1.0V/A. The DC loop gain is: AVDC = 2000 VREF RLOAD / VOUT
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Digital Camera Step-Down Power Supply
To achieve a stable circuit for the core converter, use the following procedure: 1) Determine the desired crossover frequency, either 1/3 of the zero due to the output capacitor ESR: fC = ZO 1 = 3 6 COUT ESR Discontinuous Inductor Current For discontinuous inductor current, the PWM controller has a single pole. The pole frequency and DC gain of the PWM controller are dependent on the operating duty cycle, which is: D = (2 L fOSC / RE)1/2 where RE is the equivalent load resistance, or: RE = VIN2 RLOAD / (VOUT (VOUT - VIN)) The frequency of single pole due to the PWM converter is: PO = (2 VOUT - VIN) / (2 (VOUT - VIN) RLOAD COUT) and the DC gain of the PWM controller is: AVO = 2 VOUT (VOUT - VIN) RLOAD / ((2 VOUT - VIN) D) Note that, as in the current-mode, step-down cases above, as R LOAD is increased, the pole frequency decreases and the DC gain increases proportionally. Since the crossover frequency is the product of the pole frequency and the DC gain, it remains independent of the load. As in the cases of the main and core converters, the gain through the voltage-divider is AVDV = VREF / VOUT, and the DC gain of the error amplifier is AVEA = 2000V/V. Thus, the DC loop gain is AVDC = AVDV AVEA AVO. The compensation resistor-capacitor pair at COMP cause a pole and zero at frequencies (in Hz): PC = GEA / (4000 CC) = 1 / (4x107 CC) ZC = 1 / (2 RC CC) and the ESR of the output filter capacitor causes a zero in the loop response at the frequency (in Hz): ZO = 1 / (2 COUT ESR). The DC gain and the poles and zeros are shown in the Bode plot of Figure 7. To achieve a stable circuit with the Bode plot of Figure 7, follow the procedure below: 1) Choose the RC that is equivalent to the inverse of the transconductance of the error amplifier, 1 / RC = GEA = 100s, or RC = 10k. This sets the high-frequency voltage gain of the error amplifier to 0dB. 2) Determine the maximum output pole frequency: PO(MAX) = 2VOUT - VIN 2 (VOUT - VIN )RLOAD(MIN) COUT
MAX1802
or 1/5 of the switching frequency: f fC = SW 5 whichever is lower. 2) Determine the pole frequency due to the output capacitor and the load resistor: PO = or PO = ILOAD(MAX) 2 VOUT COUT 1 2 RLOAD(MIN) COUT
3) Determine the compensation resistor required to set the desired crossover frequency: RC = 20M fC A VDC PO
or, by simplifying and using the typical VREF = 1.25V: RC = 50k/V VOUT COUT fC 4) Determine the compensation capacitor to set the proper error-amplifier pole and zero determined from the above equations: CC = 1 2 RC PO
Auxiliary Controllers The auxiliary controllers use voltage mode to regulate their output voltages. The following explains how to compensate the control system for optimal performance. The compensation differs depending on whether the inductor current is continuous or discontinuous.
where RLOAD(MIN) = VOUT / IOUT(MAX).
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23
Digital Camera Step-Down Power Supply MAX1802
3) Place the compensation zero at the same frequency as the maximum output pole frequency (in Hz): ZC = 1 2VOUT - VIN = 2RC CC 2 (VOUT - VIN )RLOAD(MIN) COUT
80 AVDC 60 PC PHASE 40 GAIN (dB) 20 GAIN O 0 Z0 FREQUENCY ZC = PO 90 PHASE 180
Solving for CC: VOUT - VIN CC = COUT VOUT RC IOUT(MAX) (2VOUT - VIN ) Use values of CC <10nF. If the above calculation determines that the capacitor should be >10nF, use CC = 10nF, skip step 4, and go to step 5. 4) Determine the crossover frequency (in Hz): VREF fC = DCOUT and to maintain at least 10dB gain margin, make sure that the crossover frequency is 1/3 of the ESR zero frequency, or 3fC ZO, or ESR D / 6 VREF. If this is not the case, go to step 5 to reduce the erroramplifier high-frequency gain to decrease the crossover frequency. 5) The high-frequency gain may be reduced, thus reducing the crossover frequency, as long as the zero due to the compensation network remains at or below the crossover frequency. In this case: ESR and fC = GEA RC VREF DCOUT 1 2 RC CC D GEA RC 6VREF
-20
Figure 7. Discontinuous-Current, Voltage-Mode, Step-Up Controller Bode Plot
increasing the phase margin. If a low-value, low-ESR output capacitor (such as a ceramic capacitor) is used, the ESR-related zero occurs at too high a frequency and does not increase the phase margin. In this case, use a lower value inductor so that it operates with discontinuous current (see the Discontinuous Inductor Current section). For continuous inductor current, the gain of the voltage divider is AVDV = VREF / VOUT, and the DC gain of the error amplifier is AVEA = 2000. The gain through the PWM controller in continuous current is: A VO = VOUT 2 VIN VREF
Thus, the total DC loop gain is: AVDC = 2000 VOUT / VIN. The complex pole pair due to the inductor and output capacitor occurs at the frequency (in Hz): PO = VOUT 2VIN LCOUT
Choose COUT, RC, and CC to satisfy both equations simultaneously. Continuous Inductor Current For continuous inductor current, there are two conditions that change, requiring different compensation. The response of the control loop includes a right-halfplane zero and a complex pole pair due to the inductor and output capacitor. For stable operation, the controller-loop gain must drop below unity (0dB) at a much lower frequency than the right-half-plane zero frequency. The zero arising from the ESR of the output capacitor is typically used to compensate the control circuit by increasing the phase near the crossover frequency,
24
The pole and zero due to the compensation network at COMP occur at the frequencies (in Hz): PC = 1 GEA = 4000CC ) 4 x 107 CC ( ZC = 1 2RC CC
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Digital Camera Step-Down Power Supply
40 AVDC 30 PC GAIN PHASE ZC = PO 90 PHASE 10 PHASE MARGIN Z0 GAIN MARGIN ZRHP FREQUENCY 180
4) Since response is 2nd order (-40dB per decade) between the complex pole pair and the ESR zero, determine the desired amplitude at the complex pole pair to force the crossover frequency equal to the ESR zero frequency. Thus: A(PO ) = (Z O / PO ) =
2
MAX1802
20 GAIN (dB)
L VIN2 COUT ESR2 VOUT 2
0
0
-10
5) Determine the desired compensation pole. Since the response between the compensation pole and the complex pole pair is 1st order (-20dB per decade), the ratio of the frequencies is equal to the ratio of the amplitudes at those frequencies. Thus: PO A = DC PC A(PO ) Solving this equation for CC: CC = VOUT (COUT )
3/2
Figure 8. Continuous-Current, Voltage-Mode, Step-Up Converter Bode Plot
The frequency (in Hz) of the zero due to the ESR of the output capacitor is: ZO = 1 2COUT ESR
ESR2
1/ 2
20M VIN (L)
and the right-half-plane zero frequency (in Hz) is: ZRHP =
(1-D)2 RLOAD
2L
6) Determine RC for the compensation zero frequency as equal to the complex pole-pair frequency: ZC = PO. Solving for RC: RC = VIN LCOUT VOUT CC
Figure 8 shows the Bode plot of the loop gain of this control circuit. To configure the compensation network for a stable control loop, set the crossover frequency at that of the zero due to the output capacitor ESR. Use the following procedure: 1) Determine the frequency of the right-half-plane zero: ZRHP
Applications Information
Using the MAX1801 with the MAX1802 Step-Down Master
The MAX1801 is a slave DC-DC controller that can be used with the MAX1802 to generate additional output voltages. The MAX1801 does not generate its own reference or oscillator. Instead it uses the reference and oscillator from the MAX1802 step-down master converter controller (Figure 1). MAX1801 controller operation and design is similar to that of the MAX1802 auxiliary controllers. For more details, refer to the MAX1801 data sheet.
(1-D)2 RLOAD =
2L 2000VOUT VIN
2) Find the DC loop gain: A VDC =
3) Determine the frequency of the complex pole pair due to the inductor and output capacitor: VOUT fO = 2 VIN LCOUT
Using an Auxiliary Controller in an SEPIC Configuration
Where the battery voltage may be above or below the required output voltage, neither a step-up nor a stepdown converter is suitable; instead, use a step-up/stepdown converter. One type of step-up/step-down
25
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Digital Camera Step-Down Power Supply
converter is the SEPIC, shown in Figure 9. Inductors L1 and L2 can be separate inductors or can be wound on a single core and coupled like a transformer. Typically, using a coupled inductor will improve efficiency since some power is transferred through the coupling, so less power passes through the coupling capacitor (C2). Likewise, C2 should have low ESR to improve efficiency. The ripple current rating must be greater than the larger of the input and output currents. The MOSFET (Q1) drain-source voltage rating and the rectifier (D1) reverse-voltage rating must exceed the sum of the input and output voltages. Other types of step-up/stepdown circuits are a flyback converter and a step-up converter followed by a linear regulator.
MAX1802
INPUT 1 CELL Li+ L1 MAIN OUTPUT 3.3V L2
ON
DCON EXT
C2 Q1
D1
R1 MAX1802 FB COMP RC GC R2
Using an Auxiliary Controller for a Multi-Output Flyback Circuit
Some applications require multiple voltages from a single converter that features a flyback transformer. Figure 10 shows a MAX1802 auxiliary controller in a two-output flyback configuration. The controller drives an external MOSFET that switches the transformer primary, and the two secondaries generate the outputs. Only a single positive output voltage can be regulated using the feedback resistive voltage-divider, so the other voltages are set by the turns ratio of the transformer secondaries. The regulation of the other secondary voltages degrades due to transformer leakage inductance and winding resistance. Voltage regulation is best when the load current is limited to a small range. Consult the transformer manufacturer for the proper design for a given application.
Figure 9. Auxiliary Controller, SEPIC Configuration
Using a Charge Pump for Negative Output Voltages
Negative output voltages can be produced without a transformer using a charge-pump circuit with an auxiliary controller as shown in Figure 11. When MOSFET Q1 turns off, the voltage at its drain rises to supply current to VOUT+. At the same time, C1 charges to the voltage at VOUT+ through D1. When the MOSFET turns on, C1 discharges through D3, thereby charging C3 to VOUT- minus the drop across D3 to create roughly the same voltage as V OUT+ at V OUT- but with inverted polarity. If different magnitudes are required for the positive and negative voltages, a linear regulator can be used at one of the outputs to achieve the desired voltage.
Conductors carrying discontinuous currents should be kept as short as possible. Conductors carrying high currents should be made as wide as possible. A separate low-noise ground plane containing the reference and signal grounds should only connect to the powerground plane at one point to minimize the effects of power-ground currents. Keep the voltage feedback network very close to the IC, preferably within 0.2in (5mm) of the FB_ pin. Nodes with high dv/dt (switching nodes) should be kept as small as possible and should stay away from highimpedance nodes such as FB_ and COMP_. Refer to the MAX1802EVKIT evaluation kit manual for a full PC board example.
Chip Information
TRANSISTOR COUNT: 7740
Designing a PC Board
A good PC board layout is important to achieve optimal performance from the MAX1802. Good design reduces excessive conducted and/or radiated noise, both of which are undesirable.
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Digital Camera Step-Down Power Supply MAX1802
INPUT 1 CELL Li+ + OUTPUT INPUT 1 CELL Li+ L Main MAIN ON DCON EXT Q1 R1 MAX1802 FB COMP COMP RC GC GC R2 RC R2 MAX1802 FB Q1 C2 R1 C1 D22 VOUT+ - OUTPUT ON DCON EXT D3 VOUTC3 D1
Figure 10. Auxiliary Controller, Flyback Configuration
Figure 11. Auxiliary Controller, Charge-Pump Configuration
Pin Configuration
COMPC
TOP VIEW
GND FBC REF
VDDC
32
31
30
29
28
27
26
FBM COMPM ONM VH VDDM DHM LXM DLM
1 2 3 4
5 6
PGND
25
LXC
VL
24 ONC 23 DCON3 22 FB3 21 COMP3
MAX1802
20 DL3 19 DL2 18 DCON2 17 COMP2
7
8
9 PGNDM
10 OSC
11 DCON1
12 13 DL1 ON1
14 COMP1
15 FB1
16 FB2
TQFP
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Digital Camera Step-Down Power Supply MAX1802
Package Information
32L TQFP, 5x5x01.0.EPS
Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are implied. Maxim reserves the right to change the circuitry and specifications without notice at any time.
28 ____________________Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600 (c) 2000 Maxim Integrated Products Printed USA is a registered trademark of Maxim Integrated Products.

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